Synchronous modulator and methods for synchronous modulation

ABSTRACT

A synchronous detector has first and second mixer circuits and a voltage controlled oscillator. The voltage controlled oscillator provides a local oscillator signal directly to the second mixer circuit and indirectly to the first mixer circuit through a phase transformer. The output of the first and second mixer circuits are combined in combiner circuitry to produce a jitter cancelled output signal. The jitter cancelled output signal is filtered in a loop filter and applied to the voltage controlled oscillator to control the frequency and phase of the local oscillator signal. The combiner circuitry includes a summer and a jitter cancellation filter. The jitter cancellation filter is preferably a high pass filter matched to spectrum of the signal detected. The output of the first mixer circuit is passed through the high pass filter into one input of the summer while the output of the second mixer circuit is passed to the second input of the summer. The output of the summer is passed to the loop filter.

This application is a division of application Ser. No. 08/223,223, filedApr. 5, 1994, pending.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The field of the invention relates to synchronous detection of RFsignals. In particular, the invention relates to synchronous detectorsproviding low phase jitter in the detected signal.

2. Description of Related Art

Known synchronous detectors use phase-locked loops to regenerate areplica of the carrier signal for use in synchronous detection. Phasemodulation of the pilot carrier is an inherent part of a vestigialsideband signal. When a phase-locked loop is used to synchronouslydetect a vestigial sideband signal, the phase-locked loop will lock ontothe inherent phase modulation of the pilot and produce replica of thecarrier signal having inherent phase noise induced thereon. This phasenoise will result in a distorted output at the output of the synchronousdemodulator. To minimize this phase noise, known phase-locked loops useloop filters with a narrow bandwidth. This limits the phase-lockedloop's ability to track phase and maintain phase coherence with thecarrier signal when sudden phase shifts are injected into the modulatedsignal, for example, undesired phase modulation in the local oscillatorof the tuner due to microphonic effects.

Synchronous detector technology is an important technology for highdefinition television, among other uses. The transmission techniques forhigh definition television is a subject of recent controversy. Someproponents desire QAM while other proponents desire VSR. For example,16-QAM, quadrature amplitude modulation, is a technique where a sequenceof four bit data nibbles are split into two separated sequences of twobit symbols per sequence. The two separated sequences of symbols are fedinto the two modulation input ports of a quadrature multiplex typemodulator. QAM output signals are double sideband signals where thesidebands bear no particular phase relationship to each other due to theasymmetry between the two separate sequences of symbols used in themodulation process.

In contrast, for example, 4-VSB, vestigial sideband, is a techniquewhere the same sequence of four bit nibbles is constituted as a singlesequence of four bit symbols where the VSB symbol rate is equal to thesum of the symbol rates of the two separated sequences of symbols usedin QAM.

The digital high definition television Grand Alliance, including AT&T,Zenith, General Instrument Corp., the Massachusetts Institute ofTechnology, Thomson Consumer Electronics, Philips Consumer Electronicsand the David Sarnoff Research Center, has selected VSB over QAM as thetransmission technology for high definition television.

The importance of high performance synchronous detection of VSB or QAMsignals to high definition television is obvious. However, the presentinvention has application to any transmission technology where unwantedphase modulation in the transmitted signal induces phase errors in thedetected signal.

SUMMARY OF THE INVENTION

It is an object of the present invention to overcome noted limitationsin the prior art. It is another object of the present invention tocancel phase noise within the bandwidth of the phase-locked loop used inthe synchronous detector. It is yet another object of the invention toimprove phase tracking accuracy for any specified loop bandwidth. It isyet another object of the invention to increase loop bandwidth inphase-locked loops for any specified phase tracking accuracy.

These and other objects are achieved in a synchronous detector havingfirst and second mixing circuits and a voltage-controlled oscillator.The voltage-controlled oscillator provides a local oscillator signal tothe second mixer circuit directly and to the first mixer circuitindirectly through a phase transformer. The output of the first andsecond mixer circuits are provided to a combiner circuit to produce acombined output signal. The combined output signal is filtered through aloop filter to provide the control signal for controlling the frequencyof the voltage controlled oscillator. The combiner circuitry included ajitter cancellation filter characterized by a transfer function having achange in signal density per unit frequency slope substantially equal toa change in signal density per unit frequency slope of the modulatedsignal to be detected.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be described in detail in the following descriptionof preferred embodiments with reference to the following figureswherein:

FIG. 1A is a graph showing a signal frequency density spectrum of aninformation signal;

FIG. 1B is a graph showing a signal frequency density spectrum of anarrow band pilot signal;

FIG. 1C is a graph showing a signal frequency density spectrum of aninformation signal containing a pilot signal;

FIG. 2 is a block diagram showing a representative system for generatingvestigial sideband modulated signals;

FIG. 3 is a block diagram showing an idealized conventional synchronousdetector;

FIGS. 4A-4D are graphs showing signal frequency density spectra ofsignals processed through the synchronous detector of FIG. 3;

FIG. 5 is a graph showing signal frequency density spectrum propertiesof the vestigial sideband modulated signal generated in the circuit ofFIG. 2;

FIGS. 6A-6D are phasor vector graphs showing the phase modulationinherent in the vestigial sideband modulated signal whose frequencydensity distribution is shown in FIG. 5;

FIG. 7 is a block diagram showing a realization of a synchronousmodulator;

FIG. 8 is a block diagram showing a finite impulse response filter;

FIG. 9 is a graph showing the impulse response of a Nyquist filter;

FIG. 10 is a z-domain plot of zeros of an order 32 realization of theNyquist filter;

FIG. 11 is a graph showing the frequency response (log magnitude) of theNyquist filter;

FIG. 12 is a graph showing the frequency response (linear amplitude) ofthe Nyquist filter;

FIG. 13 is a graph showing the impulse response of the real arm of a VSBfilter;

FIG. 14 is a graph showing the impulse response of the imaginary arm ofthe VSB filter;

FIG. 15 is a z-domain plot of the zeros of an order 32 realization ofthe VSB filter;

FIG. 16 is a graph showing the frequency response (log magnitude) of theVSB filter;

FIG. 17 is a graph showing the frequency response (linear amplitude) ofthe VSB filter;

FIG. 18 is a graph showing the frequency response of the real arm of theVSB filter;

FIG. 19 is a graph showing the phase response of the real arm of the VSBfilter;

FIG. 20 is a graph showing the magnitude response of the imaginary armof the VSB filter;

FIG. 21 is a graph showing the phase response of the imaginary arm ofthe VSB filter;

FIG. 22 is a graph showing an enlarged view of the frequency response ofthe graph of FIG. 20;

FIG. 23 is a block diagram showing a realization of a conventionalsynchronous detector;

FIG. 24 is a block diagram showing a synchronous detector according tothe present invention;

FIG. 25 is a graph showing the frequency response of the imaginary armof the VSB filter (curve A) and superimposed thereon the frequencyresponse of a first order Butterworth high-pass filter (curve B);

FIG. 26 is a graph showing the magnitude response of the first orderButterworth high-pass filter;

FIG. 27 is a graph showing the phase response of the first orderButterworth high-pass filter;

FIG. 28 is a graph showing the power spectrum at the input to loopfilter 214 of FIG. 24 using a high-pass filter (curve B) and withoutusing the high-pass filter (curve A);

FIG. 29 is a graph showing an enlarged view of FIG. 28;

FIG. 30 is a graph showing the equivalent spectrum seen by the modifiedpilot tracking circuitry. (This spectrum is not present in the modulatoror demodulator.)

FIG. 31 is a graph showing an enlarged view of the graph shown in FIG.30;

FIGS. 32A-32D are graphs and histograms showing phase tracking resultsof a simulation of the synchronous detector; and

FIG. 33 is a modulator of a transmitter incorporating a cancellationfilter according to the present invention.

DESCRIPTION OF PREFERRED EMBODIMENTS

FIG. 1A is a graph showing a signal frequency density spectrum of aninformation signal, the information signal shown being a vestigialsideband modulated signal modulated about a carrier frequency f_(c).Such a signal is characterized as having frequency components atfrequencies which are both higher and lower than the carrier frequency.However, FIG. 1A depicts a situation where the carrier signal is absent,the only frequency component at the carrier frequency being part of theinformation signal.

FIG. 1B is a graph showing a signal frequency density spectrum of apilot signal to be added to the vestigial sideband signal.

FIG. 1C is a graph showing a signal frequency density spectrum of acombination of the information signal with the pilot signal.

FIG. 2 is a block diagram showing a representative modulator forproducing a vestigial sideband signal. The signal to be modulated, m(t),whose Fourier transform is M(2πf), is first added at summation circuit 2to V pilot, where V pilot may be a dc voltage, and then mixed,preferably in a balanced mixer 4, with a carrier signal cos(2πf_(c) t)to produce a double sideband modulated signal. The double sidebandmodulated signal is processed through an appropriate filter havingtransfer function H_(v) (2πf) to produce vestigial sideband modulatedsignal φ_(v) (t) whose Fourier transform is Φ_(v) (2πf). The signalfrequency density spectrum (excluding the pilot) of the modulated signalis given by:

    Φ(2πf)=0.5[M(2πf+2πf.sub.c)+M(2πf-2πf.sub.c)]H(2πf).(1)

The original signal m(t) is recovered when the vestigial sidebandmodulated signal φ_(v) (t) is processed through a synchronous detector.

FIG. 3 is a block diagram showing a representative idealized synchronousdetector. The vestigial sideband modulated signal 100 _(v) (t) may bebandpass filtered in filter 6 and then is multiplied by the carriersignal cos(2πf_(c) t) in mixer 8 to produce a mixed signal e(t) whoseFourier transform, E(2πf), is given by:

    0.25{H(2πf+2πf.sub.c) [M(2f+4πf.sub.c)+M(2πf)]+H(2πf-2πf.sub.c) [M(2πf-4πf.sub.c)+M(2πf)]}.                      (2)

From this signal it may be necessary to remove V pilot added attransmitter (FIG. 2) using adder 9. The signal frequency densityspectrum Φ_(v) (2πf) of the vestigial sideband modulated signal is shownin FIG. 4A. The output E(2πf) of the mixer of the synchronous detectorshown in FIG. 3 is shown as a signal frequency density spectrum in FIG.4D. The first line of equation (2), above, corresponds to the signalfrequency density spectrum shown in FIG. 4C and the second line ofequation (2), above, corresponds to the signal frequency densityspectrum shown in FIG. 4B. The output of the mixer, whose Fouriertransform is E(2πf), is next processed through a low pass filter whosetransfer function is shown as dashed lines in FIG. 4D. This removes thehigh frequency content of the signal which may be present at twice thecarrier frequency. The signal frequency density spectrum of the outputof the low pass filter is given by:

    E.sub.O (2πf)=0.25M(2πf) [H(2πf+2πf.sub.c)+H(2πf-2πf.sub.c)].          (3)

The output of the low pass filter e_(O) (t) is an exact replica of themodulation signal m(t) when:

    E.sub.O (2πf)=k M(2πf),                              (4)

where k is a constant. Therefore, when the transfer function of thefilter shown in FIG. 2 is characterized by:

    H(2πf+2πf.sub.c)+H(2πf-2πf.sub.c)=4k,          (5)

then the modulation signal m(t) may be completely recovered.

The condition indicated in equation (5) is best illustrated in FIG. 5.FIG. 5 is a graph showing a signal frequency density spectrum of thevestigial sideband modulated carrier where the carrier frequency isf_(c). As can be seen from FIG. 5, point 1 is anti-symmetric to point 2with respect to the carrier frequency f_(c). Using the frequency densityof the signal frequency density spectrum at the carrier frequency f_(c)as a reference density, the frequency density at point 1 is reduced withrespect to the reference density by y₁ while the frequency density atpoint 2 is increased with respect to the reference density by y₂ .Furthermore, y₁ equals y₂ when the frequency offset at point 1 (i.e.,x₁) equals the frequency offset at point 2 (i.e., x2). A signal having afrequency spectrum as shown in FIG. 5 can be demodulated by an idealsynchronous detector to completely recover the modulation signal m(t).

Vestigial sideband modulated signals contain inherent phase modulation.FIG. 6A is a graph showing a phasor vector diagram of the carriersignal. The vector itself rotates counterclockwise in a complex voltageplane. Only the real component of the carrier signal is actually presentin the modulation process. The vector rotates counterclockwise in thecomplex voltage plane at a radian frequency of 2πf_(c) so that the realcomponent of the vector is cos(2πf_(c) t). FIG. 6B is a graph showingthe phasor vector for the upper sideband signal with respect to thecarrier signal. The phasor vector rotates counterclockwise indicating apositive frequency variance with respect to the carrier frequency. FIG.6C is a graph showing the phasor vector for the lower sideband signalwith respect to the carrier signal. The phasor vector rotates clockwiseindicating a negative frequency variance with respect to the carrierfrequency. Note that the amplitude of the phasor vector for the lowersideband is less than the amplitude of the phasor vector for the uppersideband since this signal is representative of a vestigial sidebandmodulated signal. A double sideband modulated carrier would have equalamplitude vectors, one vector for each sideband, which counter-rotateand always produce a real composite vector. However, a vestigialsideband modulated signal produces a phase modulation since the twosideband phasor vectors are unequal. It should be noted here that singlesideband modulated signals produced by filtering the output of a balancemixer produces phase modulation. In fact, since infinitely steep cutoffslopes are not possible in real filters, a signal sideband signalproduced by filtering would necessarily have a vestige sideband.

Single sideband signals produced by known methods referred to as phasingmethods also have some vestige sideband component due to limitations inrealization of wide band Hilbert filters. For example, when an audiosignal (i.e., 30 to 3840 Hz) is modulated on a carrier using a phasingmethod, the Hilbert filter is required to produce a 90° phase shift ateach frequency within the audio band (e.g., over seven octaves).

FIG. 6D is a graph showing the upper sideband phasor vector rotatingcounterclockwise as in FIG. 6B. FIG. 6D also shows, superimposed at theend of the upper sideband phasor vector, a lower sideband phasor vectorcorresponding to the phasor vector shown in FIG. 6C. There is a residualphase modulation ⊖ that remains when the upper and lower sidebands aresuperimposed. This phase modulation is inherent in the informationsignal shown in FIG. 1A, and it tends to introduce distortion into thedemodulation process in real realizations of synchronous detectors sincethe real synchronous detectors rely on a pilot signal to regenerate asynchronous replica of the carrier signal.

In U.S. Pat. No. 4,602,287, Pieter Fockens describes a system using aSAW filter to flatten the spectral power of a VSB signal near thecarrier frequency to remove the effects of this unwanted residual phasemodulation (i.e., phase jitter). This signal with the flattened spectrumis then detected by a convention synchronous detector.

A digital implementation of a vestigial sideband modulator is shown inFIG. 7. In a QAM embodiment, FIG. 7 would be modified to show two inputsrespectively feeding the imaginary and real arms. In FIG. 7 input datais applied to data input terminal 12 which is turn is provided to aninput of real arm 16 of the filter and also to an input of imaginary arm18 of the filter. Oscillator 20 generates a carrier signal at thecarrier frequency. The carrier signal is provided to both balancedmodulator 22 and an input of Hilbert filter 26. Since the carrierfrequency is substantially spectrally pure, filter 26 is merely a 90°phase shift filter at the carrier frequency. The output of filter 26 isprovided to balanced modulator 24. The output of real arm 16 is providedto balanced modulator 22 to modulate the carrier signal. The output ofbalanced modulator 22 is provided to a first input of summing circuit28. An output of imaginary arm 18 is provided to balanced modulator 24,and an output of balanced modulator 24 is provided to a second input ofsumming circuit 28. The output of summing circuit 28 is the modulatedcarrier signal and is provided at output terminal 14.

The modulator shown in FIG. 7 may produce single sideband modulationsignals when imaginary arm 18 is a Hilbert filter to provide a 90° phaseshift over the entire bandwidth of the passband of real arm filter 16.When summing circuit 28 adds the signals from balanced modulator 22 and24, the upper sideband signal is provided at output terminal 14. Whensumming circuit 28 subtracts the signal from balanced modulator 24 fromthe signal from balanced modulator 22, the modulated carrier provided atoutput terminal 14 is the lower sideband signal.

Modulator 10 may also produce a quadrature amplitude modulated (QAM)signal. In this case, as briefly described above, the data inputincludes two separate data streams feeding imaginary and real arms 16and 18. The first data stream is provided to real arm filter 16, and thesecond data stream is provided to imaginary arm filter 18. When bothreal and imaginary arms 16 and 18 are Nyquist filters, the modulatedcarrier signal at output terminal 14 is then a quadrature amplitudemodulated signal.

Modulator 10 may also produce vestigial sideband modulated signals.Modulator 10 will produce vestigial sideband modulated signals when theweights applied to the real and imaginary arms 16 and 18 areappropriately determined, as discussed in more detail below.

A digital implementation of the real and imaginary arms 16 and 18 of theVSB filter will now be described with reference to FIG. 8. The generalform of the transfer function of a discrete time filter in the z-domainis:

    H(z)=[b(z)]/[a(z)],                                        (6)

where b(z) and a(z) are polynomial expressions in z given by:

    b(z)=b.sub.0 z.sup.0 +b.sub.1 z.sup.-1 +b.sub.2 z.sup.-2 + (7)

    a(z)=a.sub.0 z.sup.0 +a.sub.1 z.sup.-1 +a.sub.2 z.sup.-2 + (8)

where z⁰ is the present sample, z⁻¹ is the next prior sample, and soforth. When it is desired to produce vestigial sideband modulatedsignals, positions of the poles and zeros of the transfer function ofthe VSB filter are rotated 45° with respect to positions of the polesand zeros of the transfer function of the Nyquist filter. Thetransformed transfer function is generally given by:

    H.sub.t (z)=[b.sub.t (z)]/[a.sub.t (z)],                   (9)

where the subscript t represents a transformed value, and thepolynomials b_(t) (z) and a_(t) (z) are given by:

    b.sub.t (z)=b.sub.0 z.sup.0 +b.sub.1 z.sup.-1 e.sup.(jω) +b.sub.2 z.sup.-2 e.sup.(j2ω)                                (10)

    a.sub.t (z)=a.sub.0 z.sup.0 +a.sub.t z.sup.-1 e.sup.(jω) +a.sub.2 z.sup.-2 e.sup.(j2ω)                                (11)

The exponential term e.sup.(jω) is responsible for the phase shift ofthe transformation which results in the vestigial sideband modulatedsignal.

Generally speaking the coefficients of the polynomial b(z) areresponsible for the zeros in the filter, and the coefficients of thepolynomial a(z) are responsible for the poles of the filter.

In the discrete time implementation discussed below, a finite impulseresponse (FIR) filter is used although other filter designs may be used.There are no poles; there are only zeros. The polynomial a_(t) (z) isequal to unity. In FIG. 8, a discrete time signal is provided to datainput 42 and passed down a tapped delay line comprised of delay elements46. Outputs from the taps are weighted by an appropriate coefficient(may be a complex number) in weighting elements 48. In the finiteimpulse response filter 40 shown in FIG. 8, the coefficients are b₁, b₂,b₃ . . . b_(N-1) and b_(N). These values correspond to the coefficientvalues of equation (7). The weighted tapped terms in filter 40 are thensummed in adder 50 and the added output is provided at output terminal44.

Although FIG. 8 depicts a discrete time FIR filter, it is common toimplement such discrete time filters as a digital filter where eachsampled data input is represented as a digital word (e.g., a 12 or 16bit word). In such a digital filter, the delay elements 46 may take theforms of clock synchronous registers which are connected in series and"clocked" at an integer multiple of the symbol rate. Furthermore, insuch a digital filter, weighting elements 48 may take the form ofdigital multipliers.

In general, a digital multiplier represented by weighting element 48, isa complex multiplier for multiplying a first complex number (a+jb) by asecond complex number (c+jd) as follows:

    (a+jb)(c+jd)=(ac-bd)+j(ad+bc),                             (12)

so that the general complex multiplier requires four real multipliersand two real adders/subtractors. Note, however, that when the data inputto terminal 12 has only a real part (i.e., b=0), the multiplier issimplified. When (a +jb) represents the output of a tap of the tappeddelay line and (c+jd) represents the weight to be multiplied by thisoutput, the output of the multiplier is ac+jad since b=0 when the inputat terminal 12 has only a real part. The real part of the output of themultiplier is ac, and the imaginary part of the output of the multiplieris ad. On this basis, the general complex filter is separated into realarm 16 for processing the ac terms and imaginary arm 18 for processingthe ad terms, both arms being provided with only the real part (i.e.,"a") of the digital input data. The real parts of the weights (i.e.,"c") are applied to real arm 16, and the imaginary parts of the weights(i.e., "c") are applied to imaginary arm 18.

In the following exemplary embodiment, a Nyquist filter will bedescribed. This filter will be described as a finite impulse responsefilter such as filter 40 shown in FIG. 8 although an equivalent infiniteimpulse response filter may be used. Filter 40 is suitable for use aseither real arm 16 or imaginary arm 18 shown in FIG. 7 (i.e., alldigital values in each filter are real). The filter described is oforder 32 (i.e., 33 taps), and the data has a sampling rate of four timesthe symbol rate. For simplicity the symbol rate will be assumed to beone Hertz; however, it will be obvious that it can be scaled to anyparticular rate.

The Nyquist filter is designed to have a transfer function correspondingto the impulse response function as shown in FIG. 9. This impulseresponse has an alpha factor of 50%. That is to say, the bandwidth ofthe filter exceeds the Nyquist bandwidth by 50%. FIG. 9 shows 33 timesamples of the impulse response. These time samples are separated by anelement of time corresponding to the time delay inherent in delayelements 46 of filter 40 (i.e., a time delay corresponding to thesampling rate). The amplitude of the impulse response functioncorresponds to the weights (i.e., b₁ . . . b_(N)) required to implementthe filter whose transfer function is given by equation (6). The valuesof these coefficients for the Nyquist filter are listed in Table 1below.

                  TABLE 1                                                         ______________________________________                                        (b.sub.0 through b.sub.32)                                                    ______________________________________                                        0.0000                                                                        -0.0008                                                                       -0.0018                                                                       -0.0021                                                                       0.0001                                                                        0.0051                                                                        0.0102                                                                        0.0101                                                                        0.0000                                                                        -0.0190                                                                       -0.0361                                                                       -0.0346                                                                       -0.0001                                                                       0.0682                                                                        0.1527                                                                        0.2229                                                                        0.2501                                                                        0.2229                                                                        0.1527                                                                        0.0682                                                                        -0.0001                                                                       -0.0346                                                                       -0.0361                                                                       -0.0190                                                                       0.0000                                                                        0.0101                                                                        0.0102                                                                        0.0051                                                                        0.0001                                                                        -0.0021                                                                       -0.0018                                                                       -0.0008                                                                       0.0000                                                                        ______________________________________                                    

The zeros of this filter are plotted in the z-domain as shown in thegraph of FIG. 10. FIG. 11 is a graph showing the frequency response (aslog magnitude) of this filter. FIG. 12 is a graph showing the frequencyresponse (as linear amplitude) of the filter. Note the symmetric shapeof the frequency response about zero frequency (which becomes thecarrier frequency after the data is modulated on the carrier signal).When both the real and imaginary arms 16 and 18 of FIG. 7 are Nyquistfilters as discussed above and each arm of the filter is provided withseparate real data streams, a quadrature amplitude modulated (QAM)signal is provided on output terminal 14.

In order to provide a vestigial sideband (VSB) modulated signal, thepoles and zeros of the Nyquist filter as shown in FIG. 10 are thenrotated by 45° or π/4 compared with FIG. 15. In the example shown inFIGS. 10 and 11, the sampling frequency of the digital filter is fourtimes the symbol rate (or 4 Hz if the symbol rate is 1 Hz). A frequencytranslation of 0.5 Hz is then performed to the filter's transferfunction. This corresponds to one-eighth of the sampling frequency or360° divided by 8 which equals 45°.

Equation (10) is used to calculate the weighting coefficients to producethe VSB filter with all poles and zeros rotated 45° with respect to thezeros depicted in FIG. 10. In general, a sequence x[n] may bemultiplied, element-by-element, with a corresponding sequencey[n]=(eω)^(n) to obtain a resulting sequence whose Fourier transform isfrequency shifted by ω from the Fourier transform of the originalsequence x[n]. In this way the poles and zeros of the sequence x[n] arethereby rotated in the z-domain about the origin by angle ω. In thepresent instance the sequence x[n] corresponds to polynomial b[z] inequations (6) and (7) which is the transfer function of a filter. Whenthe coefficients b₁ . . . b_(N) are supplied from Table 1, a Nyquistfilter results whose zeros are plotted in FIG. 10. To rotate these zerosby 45°, the values in Table 1 are multiplied, element-by-element, withthe sequence y[n]=(e.sup.ω)^(a) where ω=45° or π/4. The coefficients ofthe VSB filter are listed in Table 2.

As discussed above, the real parts of the resulting coefficients areapplied to the corresponding weighting elements in real arm 16 of theVSB filter, and the imaginary parts of the coefficients are applied tothe corresponding weighting elements of imaginary arm 18 of the VSBfilter. FIGS. 13 and 14 are graphs showing the real and imaginaryimpulse responses of this VSB filter where the values of the discretepoints in time correspond to the coefficients in Table 2. FIG. 15 is az-domain plot of the zeros produced by this VSB filter defined by thecoefficients in Table 2. Note that the positions of the zeros arerotated by 45° with respect to the positions of the zeros shown in FIG.10.

                  TABLE 2                                                         ______________________________________                                        (b.sub.0 through b.sub.32 )                                                   Real Part    Imaginary Part                                                   ______________________________________                                        0.0000       0                                                                -0.0006      -0.0006i                                                         0            -0.0018i                                                         0.0015       -0.0015i                                                         -0.0001      0                                                                -0.0036      -0.0036i                                                         0            -0.0102i                                                         0.0072       -0.0072i                                                         0.0000       0                                                                -0.0134      -0.0134i                                                         0            -0.0361i                                                         0.0245       -0.0245i                                                         0.0001       0                                                                -0.0482      -0.0482i                                                         0            -0.1527i                                                         0.1576       -0.1576i                                                         0.2501       0                                                                0.1576       +0.1576i                                                         0            +0.1527i                                                         -0.0482      +0.0482i                                                         0.0001       0                                                                0.0245       +0.0245i                                                         0            +0.0361i                                                         -0.0134      +0.0134i                                                         0.0000       0                                                                0.0072       +0.0072i                                                         0            +0.0102i                                                         -0.0036      +0.0036i                                                         -0.0001      0                                                                0.0015       +0.0015i                                                         0            +0.0018i                                                         -0.0006      +0.0006i                                                         0.0000       0                                                                ______________________________________                                    

The real components and the imaginary components are the weights appliedto the separate arms of the VSB filter since data input to the VSBfilter has only real parts.

The spectral power of the signal resulting from frequency response (aslog magnitude) for the VSB filter (i.e., both arms) after modulation onthe carrier signal is depicted in FIG. 16 (shown referenced to thecarrier frequency), and the spectral power resulting from the frequencyresponse (as linear amplitude) for the VSB filter (i.e., both arms)after modulation on the carrier signal is depicted in FIG. 17 (shownreferenced to the carrier frequency).

FIGS. 18 and 20 are graphs showing the frequency response (as linearamplitude) of the real and imaginary arms of the VSB filter,respectively. FIG. 19 is a graph showing the phase response of the realarm of the VSB filter. Note that the phase response of the real arm nearthe center frequency is zero degrees. FIG. 20 is a graph showing thefrequency response of the imaginary arm of the VSB filter as having anupper portion of the spectrum (i.e., at positive frequency) and a lowerportion of the spectrum (i.e., at negative frequency). The lower portionof the spectrum is phase shifted 180° with respect to the upper portionof the spectrum. When the real and imaginary arm components of thesignal are added together in summing element 28, the lower portion ofthe spectrum tends to cancel out, and an upper vestigial sidebandmodulation is achieved. When the imaginary arm signal is subtracted fromthe real arm signal in summing circuit 28, a lower vestigial sidebandmodulation is achieved.

The response function of the imaginary arm filter is odd symmetric. Thismeans that the phase at the negative frequency is 180° relative to thephase at the positive frequency.

FIGS. 18 and 19 are graphs showing the magnitude and phase response,respectively, of the real arm of the VSB filter. FIGS. 20 and 21 aregraphs showing the magnitude and phase response, respectively, to theimaginary arm of the VSB filter. Note the odd symmetric property shownin FIG. 21. FIG. 22 is a graph showing an enlarged portion of themagnitude response of the imaginary arm of the VSB filter generated atfrequencies near the carrier frequency.

FIG. 23 is a block diagram showing conventional synchronous detector 100which uses a phase-locked loop (PLL) to regenerate the frequency of thepilot signal in the signal from a local oscillator. Conventionalsynchronous detector 100 has detector input terminal 102 (at which amodulated carrier signal is received) and detector output terminal 104(at which a demodulated output signal is provided). Detector 100includes first mixer circuit 106 and second mixer circuit 108. A firstinput from each of first and second mixer circuit 106, 108 is connectedto detector input terminal 102. An output from first mixer circuit 106is connected to detector output terminal 104. The conventionalsynchronous detector includes a phase transformer 110, sometimesreferred to as a Hilbert filter, which provides a 90° phase shift frominput to output. The output of phase transformer 110 is connected to asecond input of first mixer circuit 106, and an input to phasetransformer 110 is connected to a second input to second mixer circuit108. The conventional synchronous detector also includesvoltage-controlled oscillator 112 (VCO) having an oscillator inputconnected to an output of second mixer circuit 108. Voltage-controlledoscillator 112 has an output connected to an input to phase transformer110. Each of first and second mixer circuits 106, 108 includes a mixerelement 106M, 108M, respectively, and a mixer filter element 106F, 108F,respectively. Mixer filter element 106F functions only as a low passfilter of the type shown in FIG. 3. Mixer filter element 108F functionsas both (1) a low pass filter of the type shown in FIG. 3, and (2) as aPLL loop filter to further limit the bandwidth of the PLL comprised ofmixer circuit 108 and voltage-controlled oscillator 112.

The modulated carrier signal applied to detector input terminal 102includes an information signal (as depicted in FIG. 1A) and a pilotsignal (as depicted in 1B). The pilot signal is characterized by adiscrete spectral line. The modulated carrier signal (as depicted inFIG. 1C) includes both the pilot signal and a portion of the informationsignal.

In operation, conventional synchronous detector 100 controls thevoltage-controlled oscillator 112 to produce an output signal having afrequency substantially equal to the frequency of the pilot signal andphase approximately coherent with the phase of the pilot signal. Thiscan only be partially achieved because the pilot signal is corrupted byinformation signal within the bandwidth of the phase lock loop. However,by designing mixer filter element 108F as a narrow band low pass filterso that only signals that are substantially at zero frequency (i.e.,near direct current signal) control the frequency of thevoltage-controlled oscillator 112, only a portion of the informationsignal, the portion having frequency components near the carrierfrequency, is averaged to control the voltage-controlled oscillator.Alternatively, a separate loop filter may be incorporated in the PLL. Ascan be seen in FIG. 5, the signal density in the upper and lowersidebands at frequencies near the carrier frequency are only slightlyout of balance, introducing only modest phase jitter as long as filter108F is narrow band. However, to achieve wide band phase-locked looptracking benefits, the bandwidth of filter 108F must be increased, thusintroducing increased phase tracking jitter generating distortion intothe detector output.

FIG. 24 is a block diagram showing synchronous detector 200 of thepresent invention. Synchronous detector 200 includes detector inputterminal 202 (at which a modulated carrier signal is received) anddetector output terminal 204 (at which a demodulated output signal isprovided). Detector 200 includes first mixer circuit 206 and secondmixer circuit 208. A first input 206a, 208a from each of first andsecond mixer circuits 206, 208, respectively, is connected to detectorinput terminal 202. First mixer output 206c from first mixer 206 isconnected to detector output terminal 204. Synchronous detector 200includes phase transformer 210, sometimes referred to as a Hilbertfilter, which provides a 90° phase shift from input 210a to output 210b.The output of phase transformer 210 is connected to second input 206b offirst mixer circuit 206, and input 210a to phase transformer 210 isconnected to second input 208b to second mixer circuit 208. Synchronousdetector 200 also includes voltage-controlled oscillator (VCO) 212having an oscillator input 212a and oscillator output 212b, oscillatoroutput 212b being connected to phase transformer input 210a. Each of thefirst and second mixer circuits 206, 208 includes a mixer element 206M,208M, respectively, and a mixer filter element 206F, 208F, respectively.Mixer filter elements 206F, 208F function only as low pass filters ofthe type described with reference to FIG. 4D. There is no need to limitthe bandwidth of mixer filter element 208F to define a loop bandwidthfor a phase-locked loop, as is required in conventional synchronousdetector 100 shown in FIG. 23.

Synchronous detector 200 includes combiner circuitry 220 which has firstand second combiner inputs 220a, 220b and combiner output 220c. Firstcombiner input 220a is connected to first mixer circuit output 206c, andsecond combiner input 220b is connected to second mixer circuit output208c. Synchronous detector 200 further includes loop filter 214 havingloop filter input 214a and loop filter output 214b. Combiner output 220cis connected to loop filter input 214a, and loop filter output 214b isconnected to oscillator input 212a.

Combiner circuitry 220 includes jitter cancellation filter 224 andsummer 222. Jitter cancellation filter 224 has an input connected tocombiner input 220a and an output connected to a first input of summer222. A second input of summer 222 is connected to second combiner input220b. An output of summer 222 is connected to combiner output 220c.

Jitter cancellation filter 224 preferably includes high pass filter 228and scaling circuit 226. Scaling circuit 226 might preferably be aresistor, although it may include an amplifier of transfer function K.

Low pass filters 206F and 208F are designed to have cutoff frequencysufficient to remove only the double carrier frequency content of themixer output as shown in FIG. 4D. Voltage-controlled oscillator 212,mixer 208, summer 222 and loop filter 214 form a phase-locked loop(PLL). The loop bandwidth is defined by loop filter 214 according todesired noise-bandwidth tradeoffs since mixer filter element 208F doesnot necessarily limit signal bandwidth. Phase jitter inherent in avestigial sideband modulated signal (or even in a single sidebandsignal) as received at the detector input terminal 202 is at leastpartially cancelled in combiner circuit 220. Because of this feature,the bandwidth of loop filter 214 need not be unnecessarily reduced inorder to minimize tracking errors due to the inherent phase modulationof a vestigial sideband modulated signal.

In operation, the signal at second mixer circuit output 208c is areplica to the signal at the output of imaginary arm 18 of the VSBfilter of modulator 10 (FIG. 7). The frequency response of imaginary arm18 (as shown in FIGS. 20 and 22) is also of the same form as shown inFIG. 25 as curve A. Superimposed on this graph (at curve B) is thefrequency response of a first order Butterworth high-pass filter with acut-off frequency of 0.1 Hz and with the gain adjusted by 1/2 (i.e., -6dB) to be shown as a close match between the two response curves. FIGS.26 and 27 are graphs showing the magnitude and phase response,respectively, of this first order Butterworth high-pass filter over amore extended frequency range. However, it should be noted that therange of interest for the synchronous detector shown in FIG. 24 islimited to the bandwidth of the loop filter. Over the bandwidth of theloop filter, jitter cancellation filter 224 is a close approximation tothe frequency response shown in FIG. 20 at like frequencies. Jittercancellation filter 224 may use any filter design so long as itsfrequency response is a close match to the spectral power distributionof the signal at second mixer circuit output 208c.

When VCO 212 is tuned to the carrier frequency the magnitude of thesignal produced at second mixer circuit output 208c is shown by curve Ain FIG. 25. The magnitude is very small for frequencies near the centerfrequency and larger for frequencies further distant from the centerfrequency. Furthermore, the phase at positive frequency differencesdiffers from the phase at negative frequency differences by 180°.

The signal at first mixer circuit output 206c is a replica of the signalat the output of real arm 16 of the VSB filter of VSB modulator 10 (FIG.7). The spectral power of this replica signal is of the same form as thefrequency response of real arm 16 as shown in FIG. 18. This replicasignal is passed through jitter cancellation filter 224. The magnitudeand phase response of jitter cancellation filter 224 is shown in FIGS.26 and 27 respectively, when the jitter cancellation filter is based onthe first order Butterworth filter described above. The spectral powerof the signal at the output of jitter cancellation filter 224 is shownby curve B in FIG. 25 when jitter cancellation filter 224 is based onthe first order Butterworth filter described above.

Thus, the frequency response of the signal at the output of the jittercancellation filter 224 is a close approximation for the frequencyresponse of the signal at second mixer circuit output 208c, at least inthe lower frequency region. It should be noted that the filter need notbe limited to a first order Butterworth filter. Any order filter andother filter types may be used so long as FIG. 25 curves A and B areapproximately matched.

Summer 222 adds the signal at the output of jitter cancellation filter224 to the signal provided at second mixer circuit output 208c. Thepower spectrum of the resulting output signal of summer 222 is shown ascurve B in FIG. 28. Curve A of FIG. 28 shows the output signal powerspectrum with jitter cancellation filter disconnected (i.e., gain equalto zero). Curve A, therefore, corresponds to the frequency responseshown in FIG. 20.

However, curve B is more subtle. The magnitude and phase of the signalinput to jitter cancellation filter 224 corresponds to the responsecurves shown in FIGS. 18 and 19, respectively. The phase near the centerfrequency is zero. Such a signal is then filtered in jitter cancellationfilter 224 having magnitude and phase responses shown in FIGS. 26 and27, respectively. At frequencies far displaced from the centerfrequency, the phase of the signal at the output of jitter cancellationfilter is still zero. In contrast, the phase of the signal (curve A,FIG. 28) at second mixer circuit output 208c is either plus or minus 90°at frequencies displaced from the center frequency. This is shown inFIGS. 20 and 21 for the signal at the output of imaginary arm 18 of theVSB filter which corresponds to the signal output from second mixercircuit output 208c. Thus, there is a 90° phase difference between thesignal at the output of jitter cancellation filter 224 and the signal atsecond mixer circuit output 208c at frequencies far displaced from thecenter frequency. Since this 90° phase difference prevents signalcancellation, the magnitude of curve B (FIG. 28) remains large atfrequencies far displaced from the center frequency.

In contrast, at frequencies near the center frequency, the phase of thesignal at the output of jitter cancellation filter 224 corresponds tothe phase response of the high-pass filter as shown in FIG. 27. Atslightly positive frequencies, there is a plus 90° phase shift, and atslightly negative frequencies, there is a minus 90° phase shift. Thephase of the signal at second mixer circuit output 208c corresponds tothe phase response of the imaginary arm of the VSB filter as shown inFIG. 21. At slightly positive frequencies, there is a minus 90° phaseshift and at slightly minus frequencies, there is a plus 90° phaseshift. At frequencies near the center frequency, summer 222 cancels thetwo input signals by adding two equal magnitude signals, one having aplus 90° phase shift and the other having a minus 90° phase shift. Thus,phase jitter inherent in a VSB signal is cancelled in combiner circuitry220. However, true phase offsets in the VCO signal with respect to thephase of the carrier signal in the modulator are detected in secondmixer circuit 208 and passed through the loop filter to adjust the VCOphase.

FIG. 28 is a graph showing the power spectrum of the signal at the inputto the loop filter. Depicted at curve B is the power spectrum using thefirst order Butterworth high-pass filter with gain set to unity.Depicted at curve A is the power spectrum at the input to the loopfilter with the high pass filter disconnected (gain set to zero). FIG.29 is a graph showing an enlarged portion of FIG. 20. Note particularlythat close to the carrier frequency, the noise density in the powerspectrum is reduced by using the high-pass filter design shown in FIG.24 as compared to old art without combiner circuitry 220, thus reducingphase jitter.

FIGS. 30 and 31 illustrate the effect of the transmission system. Ifcombiner circuitry 220 were connected into the modulator (FIG. 7) sothat the output of real arm 16 was connected to the input of jittercancellation filter 224 (i.e., combiner circuit first input 220a), andthe output of imaginary arm 18 was disconnected from mixer 24 andconnected to second combiner circuitry input 220b instead, and combinercircuitry output 220c were connected to mixer 24 in place of the outputfrom imaginary arm 18, the spectral power of the signal at outputterminal 14 would be as is shown in FIGS. 30 and 31. FIG. 30 shows theequivalent spectral power (as log magnitude) of the vestigial sidebandmodulated signal referenced to baseband when the characteristics of thefirst order Butterworth filter are used in the combiner circuitry 220.Curves A-E correspond to jitter cancellation filter gains of 0, 1.5,0.75, 1.0, and 1.25, respectively. FIG. 31 is a graph showing anenlarged portion of the frequency response curve near the carrierfrequency. In order to minimize phase jitter, it is desirable that thefrequency response near the carrier frequency be as flat as possible(e.g., curve C in FIG. 31) to counter the effects causing phase jitteras discussed with reference to FIGS. 6A-6D.

Therefore, jitter cancellation filter 224 in conjunction with summer 222in combiner circuitry 220 are capable of cancelling the phase jitterinherent in vestigial sideband and other signals at frequencies near thecarrier frequency on the transmitter side as well as the receiver side.This feature permits the bandwidth of the loop filter to be made widerwhile still maintaining a specified phase tracking accuracy. It will beappreciated that the phase tracking accuracy provided by the synchronousdetector shown in FIG. 24 may be improved compared to old art whenmaintaining a specified bandwidth for the loop bandwidth filter whichmay be a requirement due for other system constraints.

The performance of the synchronous detectors shown in FIGS. 23 and 24were simulated on a computer for a 21.5 Mbit/sec. data stream in a 4-VSBformat. FIGS. 32A-32D, are graphs and histograms showing the simulationresults for carrier recovery of 4,000 simulated symbols. The verticalscale of FIGS. 32A and 32B shows the phase of the recovered carrier ofloop filter 214 (designed as a 13.4 kHz, and N=2 low pass filter). FIG.32A corresponds to the conventional synchronous detector where the loopfilter output, when converted into degrees, varies between -1.2877 and+1.4572 degrees. The cancellation was approximated by a high pass filterwith a cut off frequency of 312.5 kHz with a mean and standard deviationof 0.0003 and 0.4629 degrees. FIG. 32B shows the loop filter output witha jitter cancellation filter (designed as the 312.5 kHz high passfilter). The excess filter bandwidth (i.e., the alpha factor) is, inthis example, 12%. The loop filter output, when converted into degrees,varies from a -0.2420 to +0.2873 degrees with a mean and standarddeviation of 0.0021 and 0.0735. Thus, synchronous detector 200 shown inFIG. 24 reduces peak-to-peak phase litter as compared to conventionalsynchronous detector 100 by a factor of more than 6 to 1 (see FIGS. 32Cand 32D for histograms of the phase of the recovered carrier). Inpractical receiver systems, the improvement enables reliable receptionof signals with less distortion (e.g., distortion known in thetelevision arts as quadrature distortion). A different filter designwhich would result in a closer match between curves A and B in FIG. 25would further improve performance. More complicated, and therefore moreexpensive, filter designs may provide better performance but mightincrease the cost and complexity of the synchronous detector. It is leftto the designer to select the filter design for the jitter cancellationfilter which provides the best phase tracking performance consistentwith the design constraints of cost, complexity and other constraintsimposed by the particular technology being used in the synchronousdetector (e.g., digital vs. analog circuits, bipolar vs. MOStransistors, etc.).

In FIG. 33 modulator 300 includes real and imaginary arm filters 16 and18 and transmitter cancellation circuit 310. Transmitter cancellationcircuit 310 corresponding to combiner circuitry 220 (FIG. 24) includeshigh pass (or band pass) filter 312, amplitude sending circuit 313, andsummer 314. The output of summer 314 provides the signal input to secondmixer 318 and real arm 16 provides the corresponding signal input tofirst mixer 316. Modulator 300 may substitute for the modulator depictedin FIG. 7 to control the shape of the spectrum as discussed withreference to FIGS. 30 and 31.

Having described preferred embodiments of a novel synchronous detectorand method for synchronous detection, which are intended to beillustrative and not limiting, it is noted that modifications andvariations can be made by those skilled in the art in light of the aboveteachings. It is therefore to be understood that changes may be made inthe particular embodiments of the invention disclosed which are withinthe scope and spirit of the invention as defined by the appended claims.

Having thus described the invention with the details and particularityrequired by the patent laws, what is desired to be protected by LettersPatent is set forth in the following claims.

What is claimed is:
 1. A modulator for shaping a spectrum of a modulatedsignal produced by modulating an information signal on a carrier signal,the modulator comprising:first and second input filters, each filtercoupled to the information signal, the first input filter having a firstfilter output, the second filter having a second filter output; combinercircuitry having first and second combiner inputs and a combiner output,the first combiner input being coupled to the first filter output, thesecond combiner input being coupled to the second filter output, thecombiner circuitry including a summer and a filter circuit, the filtercircuit having an input connected to the first combiner input, thesummer having a first summer input coupled to an output of the filtercircuit, the summer having a second summer input connected to the secondcombiner input, and the summer having a summer output connected to thecombiner output; first and second mixer circuits, each mixer circuithaving first and second mixer inputs and a mixer output, the first mixerinput of the first mixer circuit being coupled to the first filteroutput, the combiner output being coupled to the first mixer input ofthe second mixer circuit; a phase transformer having a transformer inputcoupled to the second mixer input of the first mixer circuit and havinga transformer output coupled to the second mixer input of the secondmixer circuit; a carrier signal input terminal connected to thetransformer input; and summation circuitry having a first summationcircuitry input, a second summation circuitry input and a summationcircuitry output, the first summation circuitry input being coupled tothe mixer output of the first mixer circuit, the second summationcircuitry input being coupled to the mixer output of the second mixercircuit, the summation circuitry output providing the modulated signal.2. The modulator of claim 1, wherein the filter circuit of the combinercircuitry includes one of a high pass filter and a bandpass filter. 3.The modulator of claim 1, wherein:a signal at the second filter outputis characterized by a second filter output signal slope, the secondfilter output signal slope being defined by a change in signal densityper unit frequency at frequencies within a predetermined bandwidth abouta frequency of the carrier signal applied to the carrier signal inputterminal; and the filter circuit of the combiner circuitry ischaracterized by a transfer function slope, the transfer function slopebeing defined by a change i transfer function spectral density per unitfrequency at the frequencies within the predetermined bandwidth aboutthe frequency of the carrier signal, the transfer function slope beingsubstantially equal to the second filter output signal slope.
 4. Themodulator of claim 1, wherein:the filter circuit of the combinercircuitry provides a filter circuit signal; and the second input filterprovides a second filtered signal, the second filtered signal beingsubstantially out of phase with the filter circuit signal at frequencieswithin a predetermined bandwidth about a frequency of the carrier signalapplied to the carrier signal input terminal.
 5. The modulator of claim4, wherein the filter circuit signal at least partially cancels thesecond filtered signal in the signal produced at the summer output atthe frequencies within the predetermined bandwidth about the frequencyof the carrier signal.
 6. The modulator of claim 1, wherein:the firstand second input filters provide respective first and second filteredsignals; the first filtered signal is further filtered in the filtercircuit of the combiner circuitry before it is processed by the summer;and the second filtered signal is not further filtered before it isprocessed in the summer.
 7. The modulator of claim 1, wherein the firstand second input filters are characterized by respective transferfunctions so that the modulated signal at the summation circuitry outputis a vestigial sideband modulated signal.
 8. The modulator of claim 1,wherein the filter circuit includes a first order Butterworth high passfilter.
 9. A modulator for shaping a spectrum of a modulated signalproduced by modulating an information signal on a carrier signal, themodulator comprising:first and second input filters, each filter coupledto the information signal, the first input filter having a first filteroutput, the second filter having a second filter output; combinercircuitry having first and second combiner inputs and a combiner output,the first combiner input being coupled to the first filter output, thesecond combiner input being coupled to the second filter output; firstand second mixer circuits, each mixer circuit having first and secondmixer inputs and a mixer output, the first mixer input of the firstmixer circuit being coupled to the first filter output, the combineroutput being coupled to the first mixer input of the second mixercircuit; a phase transformer having a transformer input coupled to thesecond mixer input of the first mixer circuit and having a transformeroutput coupled to the second mixer input of the second mixer circuit; acarrier signal input terminal connected to the transformer input; andsummation circuitry input and a summation circuitry output, a secondsummation circuitry input and a summation circuitry output, the firstsummation circuitry input being coupled to the mixer output of the firstmixer circuit, the second summation circuitry input being coupled to themixer output of the second mixer circuit, the summation circuitry outputproviding the modulated signal.
 10. The modulator of claim 9,wherein:the first and second input filters provide respective first andsecond filtered signals; and the combiner circuitry combines the firstand second filtered signals so that the first filtered signal at leastpartially cancels the second filtered signal in a signal produced at thecombiner output at frequencies within a predetermined bandwidth about afrequency of the carrier signal applied to the carrier signal inputterminal.
 11. The modulator of claim 10, wherein the combiner circuitryalters a phase of the first filtered signal to produce an altered phasesignal, the altered phase signal being substantially out of phase withthe second filtered signal at the frequencies within the predeterminedbandwidth about the frequency of the carrier signal, the summation ofthe altered phase signal and the second filtered signal producing thesignal provided at the combiner output.
 12. The modulator of claim 9,wherein the first and second input filters are characterized byrespective transfer functions so that the modulated signal at thesummation circuitry output is a vestigial sideband modulated signal.